Synchronizing nodes in an optical communications system utilizing frequency division multiplexing

ABSTRACT

Attenuation caused by dispersion in an optical fiber communications system is compensated. A number of low-speed channels is to be transmitted across an optical fiber. Each low-speed channel is allocated a different frequency band for transmission. The attenuation caused by dispersion is estimated for each of the frequency bands. The power of each low-speed channel is adjusted to compensate for the estimated attenuation. The power-adjusted low-speed channels are frequency division multiplexed together to produce an electrical high-speed channel suitable for transmission across the communications system.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.09/854,153, entitled “Channel Gain Control For An Optical CommunicationSystem Utilizing Frequency Division Multiplexing”, filed May 11, 2001now U.S. Pat. No. 7,228,077.

This application is a continuation-in-part of U.S. patent applicationSer. No. 09/569,761, “Channel Gain Control for an Optical CommunicationsSystem Utilizing Frequency Division Multiplexing,” by Laurence J. Newelland James F. Coward, filed May 12, 2000 now abandoned.

This application is also a continuation-in-part of U.S. patentapplication Ser. No. 09/816,242, “Through-timing of Data Transmittedacross an Optical Communications System Utilizing Frequency DivisionMultiplexing,” by David A. Pechner, et al., filed Mar. 23, 2001 now U.S.Pat. No. 7,154,914; which is a continuation-in-part of U.S. patentapplication Ser. No. 09/571,349, “Through-timing of Data Transmittedacross an Optical Communications System Utilizing Frequency DivisionMultiplexing,” by David A. Pechner and Laurence J. Newell, filed May 16,2000 now abandoned.

This application claims the benefit of U.S. Provisional PatentApplication Ser. No. 60/273,833, “High-Speed Optical Signal in anOptical Frequency Division Multiplexing System,” by Michael W. Rowan, etal., filed Mar. 6, 2001; U.S. Provisional Patent Application Ser. No.60/211,935, “Method and Apparatus to Mitigate Polarization ModeDispersion Effects in Optical Communication Networks Utilizing FrequencyDivision Multiplexing,” by Laurence J. Newell, et al., filed Jun. 15,2000; and U.S. Provisional Patent Application Ser. No. 60/209,020,“Optical Communications Networks Utilizing Frequency DivisionMultiplexing,” by Michael W. Rowan, et al., filed Jun. 1, 2000.

This application is related to U.S. patent application Ser. No.09/853,556, “Control Channel for an Optical Communications SystemUtilizing Frequency Division Multiplexing,” by Laurence J. Newell andDavid A. Pechner, filed on May 11, 2001 (now abandoned); and U.S. patentapplication Ser. No. 09/854,246, “Synchronizing Nodes in an OpticalCommunications System Utilizing Frequency Division Multiplexing,” byLaurence J. Newell, filed on May 11, 2001 (now abandoned).

The subject matter of all of the foregoing is incorporated herein byreference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates generally to optical fiber communications, andmore particularly, to the use of independent gain control for differentfrequency channels in an optical fiber communications systems utilizingfrequency division multiplexing.

2. Description of the Related Art

As the result of continuous advances in technology, particularly in thearea of networking, there is an increasing demand for communicationsbandwidth. For example, the growth of the Internet, home office usage,e-commerce and other broadband services is creating an ever-increasingdemand for communications bandwidth. Upcoming widespread deployment ofnew bandwidth-intensive services, such as xDSL, will only fartherintensify this demand. Moreover, as data-intensive applicationsproliferate and data rates for local area networks increase, businesseswill also demand higher speed connectivity to the wide area network(WAN) in order to support virtual private networks and high-speedInternet access. Enterprises that currently access the WAN through T1circuits will require DS-3, OC-3, or equivalent connections in the nearfuture. As a result, the networking infrastructure will be required toaccommodate greatly increased traffic.

Optical fiber is a transmission medium that is well suited to meet thisincreasing demand. Optical fiber has an inherent bandwidth which is muchgreater than metal-based conductors, such as twisted pair or coaxialcable. There is a significant installed base of optical fibers andprotocols such as the SONET protocol have been developed for thetransmission of data over optical fibers. The transmitter converts thedata to be communicated into an optical form and transmits the resultingoptical signal across the optical fiber to the receiver. The receiverrecovers the original data from the received optical signal. Recentadvances in transmitter and receiver technology have also resulted inimprovements, such as increased bandwidth utilization, lower costsystems, and more reliable service.

However, current optical fiber systems also suffer from drawbacks whichlimit their performance and/or utility. Many of these drawbacks arefrequency dependent. For example, optical fibers typically exhibitdispersion, meaning that signals at different frequencies travel atdifferent speeds along the fiber. More importantly, if a signal is madeup of components at different frequencies, the components travel atdifferent speeds along the fiber and will arrive at the receiver atdifferent times and/or with different phase shifts. As a result, thecomponents may not recombine correctly at the receiver, thus distortingor degrading the original signal. In fact, at certain frequencies, thedispersive effect may result in destructive interference at thereceiver, thus effectively preventing the transmission of signals atthese frequencies. Dispersion effects may be compensated by installingspecial devices along the fiber specifically for this purpose. However,the additional equipment results in additional cost and differentcompensators will be required for different types and lengths of fiber.

As another example, the electronics in an optical fiber system typicallywill have a transfer function which is not flat. That is, theelectronics will exhibit different gain at different frequencies. Inother applications, an electronic equalizer may be used to compensatefor these frequency-dependent gain variations in the electronics.However, in an optical fiber system, the electronics produce anelectrical signal which eventually is converted to/from an optical form.In order to take advantage of the wide bandwidth of optical fibers, theelectrical signal produced by the electronics preferably will have abandwidth matched to the wide bandwidth of the optical fiber. Hence, anyelectronic equalizer will also have to operate over a wide bandwidth,which makes equalization difficult and largely impractical.

Furthermore, as optical fiber systems become larger and more complex,there is an increasing need for efficient approaches to manage andcontrol these systems. In a common architecture for optical fibersystems, the system includes a set of interconnected nodes, with databeing transmitted from node to node. In these systems, there is commonlyalso a need for control, administrative or overhead information to betransmitted throughout the system or between nodes. Informationdescribing the overall network configuration, software updates,diagnostic information (including both point to point diagnostics aswell as system-wide diagnostics), timing data (such as might be requiredto implement a global clock if so desired) and performance metrics arejust a few examples of these types of information.

Thus, there is a need for optical communications systems which reduce oreliminate the deleterious effects caused by frequency-dependent effects,such as fiber dispersion and the non-flat transfer function ofelectronics in the system. There is further a need for systems whichsupport the efficient transmission of control and overhead information.

SUMMARY OF THE INVENTION

In accordance with the present invention, a method for compensating fordispersion effects in an optical fiber includes the following steps.Examples of dispersion effects are those which result from chromaticdispersion and/or polarization mode dispersion. Two or more low-speedchannels are received. Each low-speed channel is allocated a differentfrequency band for transmission across a communications system whichincludes the optical fiber. For each low-speed channel, the attenuationcaused by dispersion resulting from transmission of the low-speedchannel across the optical fiber in the frequency band allocated to thelow-speed channel is estimated. The power of each low-speed channel isadjusted to compensate for the estimated attenuation. The power-adjustedlow-speed channels are frequency division multiplexed together toproduce an electrical high-speed channel for transmission across thecommunications system. In one embodiment, gain which is equal inmagnitude to the estimated attenuation is applied to each low-speedchannel.

In another aspect of the invention, an optical fiber communicationssystem includes a variable gain block coupled to a FDM multiplexer. Thevariable gain block adjusts the power of each low-speed channel, asdescribed above. The FDM multiplexer combines the power-adjustedlow-speed channels into the electrical high-speed channel suitable fortransmission across the communications system.

The present invention is particularly advantageous because dispersioneffects, such as chromatic dispersion and/or polarization modedispersion, may be compensated in an optical fiber communicationssystem. This, in turn, enhances the performance of the overall system.

BRIEF DESCRIPTION OF THE DRAWING

The invention has other advantages and features which will be morereadily apparent from the following detailed description of theinvention and the appended claims, when taken in conjunction with theaccompanying drawing, in which:

FIG. 1A is a block diagram of a fiber optic communications system 100 inaccordance with the present invention;

FIG. 1B is a block diagram of another fiber optic communications system101 in accordance with the present invention;

FIG. 2 is a flow diagram illustrating operation of system 100;

FIG. 3A-3D are frequency diagrams illustrating operation of system 100;

FIG. 4A is a block diagram of a preferred embodiment of FDMdemultiplexer 225;

FIG. 4B is a block diagram of a preferred embodiment of FDM multiplexer245;

FIG. 5A is a block diagram of a preferred embodiment of low-speed outputconverter 270;

FIG. 5B is a block diagram of a preferred embodiment of low-speed inputconverter 275;

FIG. 6A is a block diagram of a preferred embodiment of demodulator 620;

FIG. 6B is a block diagram of a preferred embodiment of modulator 640;

FIG. 7A is a block diagram of a preferred embodiment of IFdown-converter 622;

FIG. 7B is a block diagram of a preferred embodiment of IF up-converter642;

FIG. 8A is a block diagram of a preferred embodiment of RFdown-converter 624;

FIG. 8B is a block diagram of a preferred embodiment of RF up-converter644;

FIG. 8C is a block diagram of another preferred embodiment of RFdown-converter 624; and

FIG. 8D is a block diagram of another preferred embodiment of RFup-converter 644;

FIG. 9A-9C are graphs of gain profiles resulting from attenuation due toimpairments in a fiber; and

FIG. 9D is a graph illustrating a gain ramp applied to a transmittedsignal.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1A is a block diagram of a fiber optic communications system 100 inaccordance with the present invention. System 100 includes a transmitter210B coupled to a receiver 210A by an optical fiber 104. Transmitter210B and receiver 210A are both based on frequency division multiplexing(FDM). Transmitter 210B includes an FDM multiplexer 245 coupled to anE/O converter 240. The FDM multiplexer 245 combines a plurality ofincoming signals 240B into a single signal using FDM techniques, and E/Oconverter 240 converts this single signal from electrical to opticalform 120. The E/O converter 240 preferably includes an optical source,such as a laser, and an optical modulator, such as a Mach Zendermodulator, which modulates the optical carrier produced by the opticalsource with an incoming electrical signal. For convenience, the incomingsignals 240B shall be referred to as low-speed channels; the singlesignal formed by FDM multiplexer 245 as an electrical high-speedchannel, and the final optical output 120 as an optical high-speedchannel.

Receiver 210A reverses the function performed by transmitter 210B,reconstructing the original channels 240B at the receiver location. Morespecifically, receiver 120 includes an O/E converter 220 coupled to anFDM demultiplexer 225. The O/E converter 220, preferably a detector suchas a high-speed PIN diode, converts the incoming optical high-speedchannel 120 from optical to electrical form. The frequency divisiondemultiplexer 225 frequency division demultiplexes the electricalhigh-speed channel into a plurality of low-speed channels 240A.

The various components in transmitter 210B and receiver 210A arecontrolled by their respective control systems 290. The control systems290 preferably also have an external port to allow external control ofthe transmitter 210B and receiver 210A. For example, an external networkmanagement system may manage a large fiber network, including a numberof transmitters 210B and receivers 210A. Alternately, a technician mayconnect a craft terminal to the external port to allow local control oftransmitter 210B or receiver 210A, as may be desirable duringtroubleshooting.

Various aspects of the invention will be illustrated using the examplesystem 100. However, the invention is not limited to this particularsystem 100. For example, FIG. 1B is a block diagram of another fiberoptic communications system 101 also in accordance with the presentinvention. System 101 includes two nodes 110A and 110B, each of whichincludes a transmitter 210B and receiver 210A. The two nodes 110 arecoupled to each other by two fibers 104A and 104B, each of which carriestraffic from one node 110 to the other 110. Fiber 104A carries trafficfrom transmitter 210B(A) to receiver 210A(B); whereas fiber 104B carriestraffic from transmitter 210B(B) to receiver 210A(A). In a preferredembodiment, the fibers 104 also carry control or other overhead signalsbetween the nodes 110. In an alternate embodiment, the nodes 110 may beconnected by a single fiber 104 which carries bidirectional traffic. Inother embodiments, the nodes 110 may contain additional functionality,such as add-drop functionality, thus allowing the nodes 110 to from morecomplex network configurations.

FIG. 2 is a flow diagram illustrating operation of system 100. At a highlevel, transmitter 210B combines low-speed channels 240B into an opticalhigh-speed channel 120 using FDM techniques (steps 318B, 316B and 314B).As part of this process, the power of each low-speed channel 240B isadjusted to compensate for estimated gain effects which the low-speedchannel 240B will experience while propagating through system 100 (steps321 and 323). The gain-compensated high-speed channel 120 is thentransmitted across fiber 104 (steps 312). Receiver 210A thendemultiplexes the received optical high-speed channel 120 into itsconstituent low-speed channels 240A (steps 314A, 316A and 318A).

In more detail, low-speed channels 240B are received 318B by transmitter210B. The FDM multiplexer 245 combines these channels into a high-speedchannel using frequency division multiplexing 316B techniques.Typically, each low-speed channel 240B is modulated on a carrierfrequency distinct from all other carrier frequencies and thesemodulated carriers are then combined to form a single electricalhigh-speed channel, typically an RF signal. E/O converter 240 converts314B the electrical high-speed channel to optical form, preferably viaan optical modulator which modulates an optical carrier with theelectrical high-speed channel. The optical high-speed channel 120 istransmitted 312B across fiber 104 to receiver 210A.

FIGS. 3A-3C are frequency diagrams illustrating the mapping of low-speedchannels 240B to optical high-speed channel 120 in system 100. Thesediagrams are based on an example in which high-speed channel 120 carries10 billion bits per second (Gbps), which is equivalent in data capacityto an OC-192 data stream. Each low-speed channel 240 is an electricalsignal which has a data rate of 155 million bits per second (Mbps) andis similar to an STS-3 signal. This allows 64 low-speed channels 240 tobe included in each high-speed channel 120. The invention, however, isnot to be limited by this example.

FIG. 3A depicts the frequency spectrum 310 of one low-speed channel 240Bafter pre-processing. As mentioned previously, each low-speed channel240B has a data rate of 155 Mbps. In this example, the low-speed channel240B has been pre-processed to produce a spectrally efficient waveform(i.e., a narrow spectrum), as will be described below. The resultingspectrum 310 has a width of approximately 72 MHz with low sidelobes.FIG. 3B is the frequency spectrum 320 of the electrical high-speedchannel produced by FDM multiplexer 245. Each of the 64 low-speedchannels 240B is allocated a different frequency band and thenfrequency-shifted to that band. The signals are combined, resulting inthe 64-lobed waveform 320. FIG. 3C illustrates the spectra 330 of theoptical high-speed channel 120. The RF waveform 320 of FIG. 3B isintensity modulated. The result is a double sideband signal with acentral optical carrier 340. Each sideband 350 has the same width as theRF waveform 320, resulting in a total bandwidth of approximately 11 GHz.

Receiver 210A reverses the functionality of transmitter 210B. Theoptical high-speed channel 120 is received 312A by the high-speedreceiver 210A. O/E converter 220 converts 314A the optical high-speedchannel 120A to an electrical high-speed channel, typically an RFsignal. This electrical high-speed channel includes a number oflow-speed channels which were combined by frequency divisionmultiplexing. FDM demultiplexer 225 frequency division demultiplexes316A the high-speed signal to recover the low-speed channels 240A, whichare then transmitted 318A to other destinations. The frequency spectrumof signals as they propagate through receiver 210A generally is thereverse of that shown in FIG. 3.

Note that each low-speed channel 240 has been allocated a differentfrequency band for transmission from transmitter 210B to receiver 210A.For example, referring again to FIG. 3, the low frequency channel 310Amay enter transmitter 210B at or near baseband. FDM multiplexer 245upshifts this channel 310A to a frequency of approximately 900 MHz. E/Oconverter 240 then intensity modulates this channel, resulting in twosidelobes 350A which are 900 MHz displaced from the optical carrier 340.Low-speed channel 310A propagates across fiber 104 at these particularfrequencies and is then downshifted accordingly by receiver 210A. Incontrast, the high frequency channel 310N is upshifted by FDMmultiplexer 245 to a frequency of approximately 5436 MHz and sidelobes350N are correspondingly displaced with respect to optical carrier 340.

In a preferred embodiment, the optical signal carries signals inaddition to the sidelobes 350 carrying the low-speed channels 330. FIG.3D is the frequency spectrum of an electrical high-speed channel whichalso includes a pilot tone 328 and a frequency band 326 used for controlor other overhead information. For convenience, frequency band 326 shallbe referred to as a control channel, although it may carry overheadinformation other than control signals or be used for purposes otherthan control.

In general, the control channel 326 provides a communications linkbetween the nodes along the same media (i.e., fiber 104) used by thedata-carrying sidelobes 350. The control channel 326 has many uses. Forexample, the control channel may be used for remote monitoring;performance metrics measured at one node may be communicated to anothernode or to a central location via the control channel. The controlchannel may also be used to send commands to each node, for example toset or alter the configuration of a node. When a node first comes onto anetwork or returns to the network after a fault, the control channel maybe used to implement part of the procedure for bringing the node ontothe network. For example, the control channel may be established beforethe data-carrying channels and may then be used to help set up thedata-carrying channels. Alternately, the control channel may also beused to establish handshaking between nodes. As a final example, infault situations, the control channel may be used to gather diagnosticinformation for fault isolation and also to aid in fault recovery.

The pilot tone 328 is used to synchronize local oscillators used in thetransmitter 210B and receiver 210A. The transmitter 210B generates areference signal at a frequency of 36 MHz and RF electronics attransmitter 210B are locked to this reference signal. Electronics alsogenerate the pilot tone 328 from the reference signal. In thisparticular case, the pilot tone 328 is at a frequency of 324 MHz, or theninth harmonic of the base frequency of 36 MHz. Conventional intensitymodulation results in double sideband modulation. The ninth harmonic isused in order to provide adequate separation between the pilot tones 328and the optical carrier in the final optical signal. At the receiver210A, the pilot tone 328 is recovered and frequency divided by nine torecover the original 36 MHz reference signal. Local oscillators atreceiver 210A are locked to the recovered reference signal and localoscillators at transmitter 210B are locked to the original referencesignal. Thus, local-oscillators at the receiver 210A and the transmitter210B are locked to each other.

In this embodiment, the control channel 326 has a width of 26 MHz and acenter frequency of 816 MHz. The control channel 326 is described inmore detail below. In this embodiment, both the control channel 326 andthe pilot tone 328 are located at frequencies lower than thedata-carrying sidelobes 310. However, this is not required. Alternateembodiments can locate the control channel(s) and pilot tone(s) atdifferent frequencies, including interspersed among the sidelobes 310and/or at frequencies higher than the sidelobes 310.

Since each low-speed channel 240 is allocated a different frequencyband, each channel will typically experience a different gain as itpropagates through system 100. For example, fiber losses, such as due tochromatic dispersion or polarization mode dispersion, typically will bedifferent for sidelobes 350A and 350N since they are located atdifferent frequencies. Similarly, the gain due to propagation throughthe various electronic components may also differ since electronics mayexhibit different responses at different frequencies. The term “gain” isused here to refer to both losses and amplification.

However, since the frequency band of each low-speed channel 240 isknown, the gain which the low-speed channel 240 will experience as itpropagates through system 100 may be estimated 323 and then compensatedfor 321 by adjusting the power of each low-speed channel. For example,if sidelobe 350N is expected to experience more loss than sidelobe 350Adue to chromatic dispersion, then sidelobe 350N may be amplified withrespect to sidelobe 350A in order to compensate for the expected higherloss. The amplification may be applied directly to sidelobe 350N or atother locations within system 100, for example to lobe 310N exiting theFDM multiplexer 245 or to the corresponding low-speed channel 240B as itenters the system 100.

The gain may be estimated in any number of ways. For example, withrespect to fiber 104, in one embodiment, standard analytical models areused to estimate the gain due to propagation through fiber 104 atdifferent frequencies due to different physical phenomena. Often, thesegain estimates will depend on the length of fiber 104, which itself maybe estimated based on the expected application. Alternately, the lengthmay be measured, for example by using time-domain reflectometry. In apreferred embodiment, a test signal is sent from node 110A over fiber104A to node 110B. Node 110B receives the signal and then returns it tonode 110A via fiber 104B A timer circuit measures the round-trip elapsedtime, which is used to estimate the fiber length.

Similarly, the gain estimates for fiber 104 may alternately bedetermined empirically by measuring the actual gain experienced atdifferent frequencies or by using empirical models. Analogous techniquesmay be applied to the rest of system 100. For example, the gain ofelectronics may be estimated based on models or may be measured bycalibrations, for example performed by the manufacturer at the time ofproduction.

FIGS. 9A-9C are graphs illustrating the attenuation resulting fromchromatic dispersion. These graphs plot gain, so increased attenuationis shown as low values of gain. Generally speaking, in optical systemsusing double-sideband optical signals, the attenuation of the detectedsignal which results from chromatic dispersion is a function of thelength of the fiber, denoted by 1, and the frequency of the sidelobe 350of interest, denoted by f. As shown in FIG. 9A, for a given frequency f,chromatic dispersion results in an increasing attenuation withincreasing length l, until a null is reached. After a null is reached,the attenuation decreases. Similarly, as shown in FIG. 9B, for a givenlength of fiber 1, the attenuation due to chromatic dispersion increaseswith increasing frequency f, until a null is reached. Then, theattenuation decreases. If the fiber length l and frequencies f of thesidelobes 350 are selected so that a null is not reached, then thechromatic dispersion typically results in a gain rolloff with frequencyin the detected signal, as shown in FIG. 9C. Polarization modedispersion generally has a similar behavior.

Thus, if all of the sidelobes 350 were of equal power when they entereda fiber 104 with the gain profile shown in FIG. 9C, the higher frequencysidelobes typically would experience more attenuation in the detectedsignal as the optical signal propagates through the fiber. This wouldresult in a rolloff in power received at the receiver 210A at the higherfrequencies. Since it is desirable for power for all sidelobes 350 to beroughly equal at the receiver 210A, it is desirable to compensate forthis rolloff effect. Accordingly, at the transmitter 210B, the power ofthe higher frequency low-speed channels 240 is boosted 321 with respectto the lower frequency channels 240 so that after propagation throughfiber 104, the sidelobes 350 are of roughly equal power when they reachthe receiver 210A. FIG. 9D is a graph of the gain G applied tocompensate for the rolloff. As the inverse of gain g in FIG. 9C (i.e.,G=1/g), the gain G in FIG. 9D increases with increasing frequency and isconcave up. This gain profile is also known as a gain ramp. The gain Gis shown as a continuous curve. However, in a preferred embodiment, aconstant gain is applied across each sidelobe 350. For example, the gainG at the center frequency of a specific sidelobe 350 may be applied tothe entire sidelobe.

When more than one effect is present, the gain G preferably compensatesfor all significant effects. For example, in some situations, bothchromatic dispersion and polarization mode dispersion result insubstantial attenuation of the signal. In one embodiment, thecompensatory gain function G(f) is determined according toG(f)=G_(CD)(f) G_(PMD)(f), where G_(CD)(F) compensates for attenuationdue to chromatic dispersion and G_(PMD)(f) compensates for attenuationdue to polarization mode dispersion. In one embodiment, the functionG_(PMD)(ƒ) is selected to accommodate for the peak instantaneousdifferential group delay intended to be tolerated. In a preferredembodiment, the gain G_(PMD)(ƒ) compensates for a peak differentialgroup delay of 46 ps and results in a 3 dB gain applied to low-speedchannel number 64, centered at frequency f=5436 MHz. This 3 dB gainoffsets the differential group delay of 46 ps and ensures that datachannel 64 arrives with the same power as a data channel propagatingwithout substantial PMD and therefore without a gain ramp. Continuingthis example, an instantaneous differential group delay of 70 ps due topolarization mode dispersion results in an optical power penalty of 3dB.

Other compensatory gain functions G will be apparent. For example, theexternal optical modulator in E/O converter 240 may result in a rolloffwith frequency. The gain G can be used to compensate for this rolloff,for example by using a power amplifier to apply gain to the RF signalentering the modulator.

The gain may also be estimated using closed loop techniques. In otherwords, the low-speed channel 240 is transmitted across system 100 and afeedback signal is produced responsive to this transmission. The powerof the low-speed channel is then adjusted 321 responsive to the feedbacksignal. As examples, in one embodiment, the feedback signal may dependon the power of the low-speed channel after it has been transmittedacross system 100. In another embodiment, it may depend on the signal tonoise ratio or various error rates in the received low-speed channel240A.

In a preferred embodiment, the feedback signal is generated by monitorcircuitry coupled to the FDM demultiplexer 225 and fed back fromreceiver 210A to transmitter 210B via fiber 104, as opposed to someother communications channel. In system 101 of FIG. 1B, the controlsystems 290 may communicate with each other via the bidirectionaltraffic on these fibers 104. For example, consider traffic flow fromtransmitter 210B(A) across fiber 104A to receiver 210A(B). The feedbacksignal generated at receiver 210A(B) for this traffic is fed back totransmitter 210B(A) via the other fiber 104B. The control system 290 fornode 110A then generates the appropriate control signals to adjust thepowers of the low-speed channels. Similarly, the feedback signal fortraffic flowing from transmitter 210B(B) across fiber 104B to receiver210A(A) may be fed back to transmitter 210B(B) via the other fiber 104A.

In a preferred embodiment, a frequency band located between thesidebands 350 (see FIG. 3C) and the optical carrier 340 is allocated forcontrol and/or administrative purposes (e.g., for downloading softwareupdates). In a preferred embodiment, this control channel is also usedto transmit the feedback signal between the nodes 110 and for timedomain reflectometry in order to estimate the length of the fiber. Sinceit is often desirable to establish initial communications between nodes110 using the control channel before establishing the actual data linksusing sidebands 350, the control channel preferably has a lower datarate and is less susceptible to transmission impairments than the datacarrying sidebands 350. In an alternate embodiment, one of the frequencybands within the electrical high-speed channel 320 is used for thefeedback signal.

Referring now to FIG. 3D, in one embodiment, the control channel 326 hasa spectral bandwidth of 26 MHz and utilizes alternate markinversion/frequency-shift keying (AMI/FSK) modulation with a peakfrequency deviation of 9 MHz. Data is transmitted at a rate of 2.048Mbps using the E1 protocol. Because the control channel 326 transmits atthe E1 data rate, which is lower than the transmission rate of thedata-carrying sidebands 310, control channel 326 is more robust than thedata channels 310 and can tolerate lower SNR. Furthermore, because ofthe lower data rate and because, in the optical signal, the controlchannel 326 is closer to the optical carrier than the data-carryingchannels 350, the control channel 326 is generally more resistant tofiber impairments than the data channels 350. Thus, in situations whenthe data channels 350 are not transmitting properly, the control channelmay still be functioning normally. The control channel 326 can then beused by control system 290 to communicate between nodes 110A and 110B inorder to bring the data channels 350 to normal operation. This situationmay occur if there is a fault in the system or upon start up of thesystem. The control channel 326 can also be used to exchange informationduring routine operation, as described above.

Any number of techniques may be used to adjust 321 the power of thelow-speed channels 240. For example, if a closed loop technique is used,standard control algorithms such as proportional control may be used. Inanother approach, a common mode and a differential mode adjustment maybe used alternately. In the differential mode adjustment, the totalpower of all low-speed channels is kept constant while the allocation ofpower among the various channels is adjusted. Thus, for example, thegain applied to sidelobe 350A may be increased by a certain amount ifthe gain applied to sidelobe 350N is reduced by the same amount, so thatthe total power in all sidelobes 350 remains constant. In the commonmode adjustment, the allocation of power among the various low-speedchannels 240 remains constant while the total power is adjusted. Thus,for example, the gain applied to sidelobes 350A, 350N and all othersidelobes 350 may be increased by the same amount, thus increasing thetotal power.

The use of frequency division multiplexing in system 100 allows thetransport of a large number of low-speed channels 240 over a singlefiber 104 in a spectrally-efficient manner. It also reduces the cost ofsystem 100 since the bulk of the processing performed by system 100 isperformed on low-speed electrical signals. In addition, since eachlow-speed channel is allocated a specific frequency band, the use offrequency division multiplexing allows different gain to be applied toeach low-speed channel in an efficient manner, thus compensating for thespecific gain to be experienced by the low-speed channel as itpropagates through system 100.

FIGS. 4-8 are more detailed block diagrams illustrating various portionsof a preferred embodiment of system 100. Each of these figures includesa part A and a part B, which correspond to the receiver 210A andtransmitter 210B, respectively. These figures will be explained byworking along the transmitter 210B from the incoming low-speed channels240B to the outgoing high-speed channel 120, first describing thecomponent in the transmitter 120B (i.e., part B of each figure) and thendescribing the corresponding components in the 120A (i.e., part A ofeach figure). These figures are based on the same example as FIG. 3,namely 64 STS-3 data rate low-speed channels 240 are multiplexed into asingle optical high-speed channel 120. However, the invention is not tobe limited by this example or to the specific structures disclosed.

FIG. 4B is a block diagram of a preferred embodiment of transmitter210B. In addition to FDM multiplexer 245 and E/O converter 240, thistransmitter 210B also includes a low-speed input converter 275 coupledto the FDM multiplexer 245. FDM multiplexer 245 includes a modulator640, IF up-converter 642, and RF up-converter 644 coupled in series.FIGS. 6B-8B show further details of each of these respective components.Similarly, FIG. 4A is a block diagram of a preferred embodiment ofreceiver 210A. In addition to O/E converter 220 and FDM demultiplexer225, this receiver 210A also includes a low-speed output converter 270coupled to the FDM demultiplexer 225. FDM demultiplexer 225 includes anRF down-converter 624, IF down-converter 622, and demodulator 620coupled in series, with FIGS. 6A-8A showing the corresponding details.

FIGS. 5A-5B are block diagrams of one type of low-speed converter270,275. In the transmit direction, low-speed input converter 275converts tributaries 160B to low-speed channels 240B, which have thesame data rate as STS-3 signals in this embodiment. The structure ofconverter 275 depends on the format of the incoming tributary 160B. Forexample, if tributary 160B is an STS-3 signal then no conversion isrequired. If it is an OC-3 signal, then converter 275 will perform anoptical to electrical conversion.

FIG. 5B is a converter 275 for an OC-12 tributary. Converter 275includes an O/E converter 510, CDR 512, TDM demultiplexer 514, andparallel to serial converter 516 coupled in series. The O/E converter510 converts the incoming OC-12 tributary 160B from optical toelectrical form, producing the corresponding STS-12 signal. CDR 512performs clock and data recovery of the STS-12 signal and alsodetermines framing for the signal. CDR 512 also converts the incomingbit stream into a byte stream. The output of CDR 512 is byte-wide, asindicated by the “×8.” Demultiplexer 514 receives the signal from CDR512 one byte at a time and byte demultiplexes the recovered STS-12signal using time division demultiplexer (TDM) techniques. The result isfour separate byte-wide signals, as indicated by the “4×8,” each ofwhich is equivalent in data rate to an STS-3 signal and with thecorresponding framing. Converter 516 also converts each byte-wide signalinto a serial signal at eight times the data rate, with the resultingoutput being four low-speed channels 240B, each at a data rate of 155Mbps.

Low-speed input converter 270 of FIG. 5A implements the reversefunctionality of converter 275, converting four 155 Mbps low-speedchannels 240A into a single outgoing OC-12 tributary 160A. Inparticular, converter 270 includes CDR 528, FIFO 526, TDM multiplexer524, parallel to serial converter 522, and E/O converter 520 coupled inseries. CDR 528 performs clock and data recovery of each of the fourincoming low-speed channels 240A, determines framing for the channels,and converts the channels from serial to byte-wide parallel. The resultis four byte-wide signals entering FIFO 526. FIFO 526 is a buffer whichis used to synchronize the four signals in preparation for combiningthem into a single STS-12 signal. Multiplexer 524 performs the actualcombination using TDM, on a byte level, to produce a single byte-widesignal equivalent in data capacity to an STS-12 signal. Parallel toserial converter 522 adds STS-12 framing to complete the STS-12 signaland converts the signal from byte-wide parallel to serial. E/O converterconverts the STS-12 signal to electrical form, producing the outgoingOC-12 tributary 160A.

Converters 270 and 275 have been described in the context of OC-3 andOC-12 tributaries and low-speed channels with the same date rate asSTS-3 signals, but the invention is not limited to these protocols.Alternate embodiments can vary the number, bit rate, format, andprotocol of some or all of these tributaries 160. One advantage of theFDM approach illustrated in system 100 is that the system architectureis generally independent of these parameters. For example, thetributaries 160 can comprise four 2.5 Gbps data streams, 16 622 Mbpsdata streams, 64 155 Mbps data streams, 192 51.84 Mbps data streams, orany other bit rate or combinations of bit rates, without requiring majorchanges to the architecture of system 100.

In one embodiment, the tributaries 160 are at data rates which are notmultiples of the STS-3 data rate. In one variant, low-speed inputconverter 275 demultiplexes the incoming tributary 160B into some numberof parallel data streams and then stuffs null data into each resultingstream such that each stream has an STS-3 data rate. For example, iftributary 160B has a data rate of 300 Mbps, converter 275 maydemultiplex the tributary into four 75 Mbps streams. Each stream is thenstuffed with null data to give four 155 Mbps low-speed channels. Inanother variant, the speed of the rest of system 100 (specifically themodulator 640 and demodulator 620 of FIG. 4) may be adjusted to matchthat of the tributary 160. Low-speed output converter 270 typically willreverse the functionality of low-speed input converter 275.

Referring to FIG. 6B, modulator 640 modulates the 64 incoming low-speedchannels 240B to produced 64 QAM-modulated channels which are input tothe IF up-converter 642. For convenience, the QAM-modulated channelsshall be referred to as IF channels because they are inputs to the IFup-converter 642. In this embodiment, each low-speed channel 240 ismodulated separately to produce a single IF channel and FIG. 6B depictsthe portion of modulator 640 which modulates one IF channel. Modulator640 in its entirety would include 64 of the portions shown in FIG. 6B.For convenience, the single channel shown in FIG. 6B shall also bereferred to as a modulator 640. Modulator 640 includes a FIFO 701,Reed-Solomon encoder 702, an interleaver 704, a trellis encoder 706, adigital filter 708 and a D/A converter 710 coupled in series. Modulator640 also includes a synchronizer 712 coupled between the incominglow-speed channel 240B and the filter 708.

Modulator 640 operates as follows. FIFO 701 buffers the incominglow-speed channel. Reed-Solomon encoder 702 encodes the low-speedchannel 240B according to a Reed-Solomon code. Programmable Reed-Solomoncodes are preferred for maintaining very low BER (typ. lower than 10⁻¹²)with low overhead (typ. less than 10%). This is particularly relevantfor optical fiber systems because they generally require low bit errorrates (BER) and any slight increase of the interference or noise levelwill cause the BER to exceed the acceptable threshold. For example, aReed-Solomon code of (204,188) can be applied for an error correctioncapability of 8 error bytes per every 204 encoded bytes.

The interleaver 704 interleaves the digital data string output by theReed-Solomon encoder 702. The interleaving results in more robust errorrecovery due to the nature of trellis encoder 706. Specifically, forwarderror correction (FEC) codes are able to correct only a limited numberof mistakes in a given block of data, but convolutional encoders such astrellis encoder 706 and the corresponding decoders tend to cause errorsto cluster together. Hence, without interleaving, a block of data whichcontained a large cluster of errors would be difficult to recover.However, with interleaving, the cluster of errors is distributed overseveral blocks of data, each of which may be recovered by use of the FECcode. Convolution interleaving of depth 0 is preferred in order tominimize latency.

The trellis encoder 706 applies a QAM modulation, preferably 16 stateQAM modulation, to the digital data stream output by the interleaver704. The result typically is a complex baseband signal, representing thein-phase and quadrature (I and Q) components of a QAM-modulated signal.Trellis encoder 706 implements the QAM modulation digitally and theresulting QAM modulated signal is digitally filtered by filter 708 inorder to reduce unwanted sidelobes and then converted to the analogdomain by D/A converter 710. Synchronizer 712 performs clock recovery onthe incoming low-speed channel 240B in order to synchronize the digitalfilter 708. The resulting IF channel is a pair of differential signals,representing the I and Q components of the QAM-modulated signal. Inalternate embodiments, the QAM modulation may be implemented usinganalog techniques.

Referring to FIG. 6A, demodulator 620 reverses the functionality ofmodulator 640, recovering a low-speed channel 240A from an incoming IFchannel (i.e., analog I and Q components in this embodiment) receivedfrom the IF down-converter 622. Demodulator 620 includes an A/Dconverter 720, digital Nyquist filter 722, equalizer 724, trellisdecoder 726, deinterleaver 728, Reed-Solomon decoder 730 and FIFO 732coupled in series. Demodulator 620 further includes a synchronizer 734which forms a loop with Nyquist filter 722 and a rate converterphase-locked loop (PLL) 736 which is coupled between synchronizer 734and FIFO 732.

Demodulator 620 operates as FIG. 6A would suggest. The A/D converter 720converts the incoming IF channel to digital form and Nyquist filter 722,synchronized by synchronizer 734, digitally filters the result to reduceunwanted artifacts from the conversion. Equalizer 724 appliesequalization to the filtered result, for example to compensate fordistortions introduced in the IF signal processing. Trellis decoder 726converts the I and Q complex signals to a digital stream anddeinterleaver 728 reverses the interleaving process. Trellis decoder 726may also determine the error rate in the decoding process, commonlyreferred to as the channel error rate, which may then be used toestimate the gain of system 100 as described previously. Reed-Solomondecoder 730 reverses the Reed-Solomon encoding, correcting any errorswhich have occurred. If the code rate used results in a data rate whichdoes not match the rate used by the low-speed channels, FIFO 732 andrate converter PLL 736 transform this rate to the proper data rate.

Referring again to transmitter 210B, IF up-converter 642 receives the 64IF channels from modulator 640. Together, IF up-converter 642 and RFup-converter 644 combine these 64 IF channels into a single RF signalusing FDM techniques. In essence, each of the IF channels (orequivalently, each of the 64 low-speed channels 240B) is allocated adifferent frequency band within the RF signal. The allocation offrequency bands shall be referred to as the frequency mapping, and, inthis embodiment, the IF channels may also be referred to as FDM channelssince they are the channels which are FDM multiplexed together. Themultiplexing is accomplished in two stages. IF up-converter 642 firstcombines the 64 IF channels into 8 RF channels, so termed because theyare inputs to the RF up-converter 644. In general, the terms “IF” and“RF” are used throughout as labels rather than, for example, indicatingsome specific frequency range. RF up-converter 644 them combines the 8RF channels into the single RF signal, also referred to as theelectrical high-speed channel.

Referring to FIG. 7B, IF up-converter 642 includes eight stages(identical in this embodiment, but not necessarily so), each of whichcombines 8 IF channels into a single RF channel. FIG. 7B depicts one ofthese stages, which for convenience shall be referred to as an IFup-converter 642. IF up-converter 642 includes eight frequency shiftersand a combiner 812. Each frequency shifter includes a modulator 804, avariable gain block 806, a filter 808, and a power monitor 810 coupledin series to an input of the combiner 812.

IF up-converter 642 operates as follows. Modulator 804 receives the IFchannel and also receives a carrier at a specific IF frequency (e.g.,1404 MHz for the top frequency shifter in FIG. 7B). Modulator 804modulates the carrier by the IF channel. The modulated carrier isadjusted in amplitude by variable gain block 806, which is controlled bythe corresponding control system 290, and bandpass filtered by filter808. Power monitor 810 monitors the power of the gain-adjusted andfiltered signal, and transmits the power measurements to control system290.

In a preferred embodiment, each IF channel has a target power levelbased on the estimated gain due to transmission through system 100.Control system 290 adjusts the gain applied by variable gain block 806so that the actual power level, as measured by power monitor 810,matches the target power level. The target power level may be determinedin any number of ways. For example, the actual power level may berequired to fall within a certain power range or be required to alwaysstay above a minimum acceptable power. Alternately, it may be selectedto maintain a minimum channel error rate or to maintain a channel errorrate within a certain range. In this embodiment, variable gain block 806implements the step of adjusting 321 the power of each low-speed channel240.

In alternate embodiments, the power adjustment may be implemented byother elements at other locations or even at more than one location. Forexample, one gain block may apply a common mode gain to all low-speedchannels, and another series of gain blocks at a different location mayapply individual gain to each channel (i.e., differential mode gain).However, applying the gain adjustment at the location of variable gainblock 806 has some advantages. For example, if the power were adjustedprior to modulator 804, where each low-speed channel consists of an Iand a Q channel, care would need to be taken to ensure that the samegain was applied to both the I and Q channels in order to preventdistortion of the signal. Alternately, if the power were adjusted aftercombiner 812, it typically would be more difficult to adjust the powerof each individual low-speed channel since combiner 812 produces acomposite signal which includes multiple individual channels.

The inputs to combiner 812 are QAM-modulated IF signal at a specificfrequency which have been power-adjusted to compensate for estimatedgains in the rest of system 100. However, each frequency shifter uses adifferent frequency (e.g., ranging in equal increments from 900 MHz to1404 MHz in this example) so combiner 812 simply combines the 8 incomingQAM-modulated signal to produce a single signal (i.e., the RF channel)containing the information of all 8 incoming IF channels. In thisexample, the resulting RF channel covers the frequency range of 864-1440MHz.

Referring to FIG. 8B, RF up-converter 644 is structured similar to IFup-converter 642 and performs a similar function combining the 8 RFchannels received from the IF up-converter 642 just as each IFup-converter combines the 8 IF channels received by it. In more detail,RF up-converter 644 includes eight frequency shifters and a combiner912. Each frequency shifter includes a mixer 904, various gain blocks906, and various filter 908 coupled in series to an input of thecombiner 912.

RF up-converter 644 operate as follows. Mixer 904 mixes one of the RFchannels with a carrier at a specific RF frequency (e.g., 4032 MHz forthe top frequency shifter in FIG. 8B), thus frequency upshifting the RFchannel to RF frequencies. Gain blocks 906 and filters 908 are used toimplement standard amplitude adjustment and frequency filtering. Forexample, in FIG. 8B, one filter 908 bandpass filters the incoming RFchannel and another bandpass filters the produced RF signal, bothfilters for suppressing artifacts outside the frequency range ofinterest. Each frequency shifter uses a different frequency (e.g.,ranging in equal increments from 0 to 4032 MHz in this example) socombiner 912 simply combines the 8 incoming RF signals to produce thesingle electrical high-speed channel containing the information of all 8incoming RF channels or, equivalently, all 64 IF channels received by IFup-converter 642. In this example, the electrical high-speed channelcovers the frequency range of 864-5472 MHz.

RF down-converter 624 and IF down-converter 622 implement the reversefunctionalities, splitting the RF signal into its 8 constituent RFchannels and then splitting each RF channel into its 8 constituent IFchannels, respectively, thus producing 64 IF channels (i.e., FDMchannels) to be received by demodulator 620.

Referring to FIG. 8A, RF down-converter 624 includes a splitter 920coupled to eight frequency shifters. Each frequency shifter includes amixer 924, various gain blocks 926, and various filters 928 coupled inseries. Splitter 920 splits the incoming electrical high-speed channelinto eight different RF signals and each frequency shifter recovers adifferent constituent RF channel from the RF signal it receives. Mixer924 mixes the received RF signal with a carrier at a specific RFfrequency (e.g., 4032 MHz for the top frequency shifter in FIG. 8A),thus frequency downshifting the RF signal to its original IF range(e.g., 864-1440 MHz). Filter 928 then filters out this specific IFfrequency range. Each frequency shifter uses a different RF frequencywith mixer 924 and thus recovers a different RF channel. The output ofRF down-converter 624 is the 8 constituent RF channels.

IF down-converter 622 of FIG. 7A operates similarly. It includes asplitter 820 and 8 frequency shifters, each including a bandpass filter822, variable gain block 823, demodulator 824, and power monitor 826.Splitter 820 splits the incoming RF channel into eight signals, fromwhich each frequency shifter will recover a different constituent IFchannel. Filter 822 isolates the frequency band within the RF channelwhich contains the IF channels of interest. Demodulator 824 recovers theIF channel by mixing with the corresponding IF carrier. The resulting 64IF channels are input to demodulator 620.

Variable gain block 823 and power monitor 826 control the power level ofthe resulting IF channel. In a preferred embodiment, each IF channel isoutput from IF down-converter 622 at a target power in order to enhanceperformance of the rest of the receiver 210A. Power monitor 826 measuresthe actual power of the IF channel, which is used to adjust the gainapplied by variable gain block 823 in order to match the actual andtarget power levels. As described previously, the actual received powerlevel for each low-speed channel may be used to estimate the gain ofsystem 100. In IF down-converter 622, the actual receive power level maybe determined by dividing the output target power for each IF channel bythe gain applied by variable gain block 823 in order to maintain theoutput target power. In another approach, the actual receive power levelmay be directly measured, for example by placing a power monitor wherevariable gain block 823 is located.

FIGS. 8C and 8D are block diagrams of the RF downconverter 624 and RFupconverter 622, respectively, which explicitly account for the pilottone 328 and control channel 326. The RF downconverter 624 in FIG. 8C isthe same as that in FIG. 8A except for the following difference. In FIG.8C, the splitter 920 splits the incoming signal into ten parts, ratherthan eight, and the RF downconverter 624 includes two additional signalpaths coupled to splitter 920 to process the two additional parts. Inthis example, each of the additional signal paths includes a filter 928coupled to a variable gain block 926. The first signal path with filter928 centered at 816 MHz recovers the control channel 326 and the secondwith filter 928 centered at 324 MHz recovers the pilot tone 328.

The RF upconverter 644 in FIG. 8D is changed in a similar manner.Specifically, in addition to the eight signal paths leading to combiner912 shown in FIG. 8C, the RF upconverter in FIG. 8D includes twoadditional signal paths. Each signal path includes a variable gain block908 coupled in series to a filter 908. One path is for adding thecontrol channel 326 and the other adds the pilot tone 328.

A preferred embodiment of method 300 will now be described, withreference to the bidirectional system 101 and the further details givenin FIGS. 5-8. In the preferred method, the gain applied to eachlow-speed channel 240 is adjusted in order to optimize the channel errorrate measured at the receiver 210A. Feedback occurs over fibers 104.More specifically, gain is applied to each of the low-speed channels 240via variable gain block 806. This gain is initially selected based on anopen-loop estimate. As data is transmitted from transmitter 210B(A) overfiber 104A to receiver 210A(B), trellis decoder 726 determines thechannel error rate at the receiver 210A(B). The channel error rate isfed back to node 110A via the control channel on fiber 104B. In thisembodiment, the control channel is a frequency modulated, alternate markinverted, B8ZS-encoded baseband transmitted at 2 Mbps. The gain appliedby variable gain block 806 is adjusted to optimize this channel errorrate. One optimization approach alternates between differential mode andcommon mode adjustments. In the differential mode adjustment, the gainis increased for low-speed channels 240 which have unacceptable channelerror rates and decreased for low-speed channels 240 with acceptablechannel error rates, while keeping the overall power in all low-speedchannels constant. In the common mode adjustment, if the median channelerror rate is unacceptable, then the gain for all channels 240 isincreased by equal increments until the median channel error rate isacceptable. In alternate embodiments, channel performance can bemonitored by metrics other than the channel error rate, for example,received power, signal to noise ratio, or bit error rate.

It should be noted that many other implementations which achieve thesame functionality as the devices in FIGS. 5-8 will be apparent. Forexample, referring to FIG. 8B, note that the bottom channel occupies thefrequency spectrum from 864-1440 MHz and, therefore, no mixer 904 isrequired. As another example, note that the next to bottom channel isfrequency up shifted from the 864-1440 MHz band to the 1440-2016 MHz. Ina preferred approach, this is not accomplished in a single step bymixing with a 576 MHz signal. Rather, the incoming 864-1440 MHz signalis frequency up shifted to a much higher frequency range and thenfrequency down shifted back to the 1440-2016 MHz range. This avoidsunwanted interference from the 1440 MHz end of the original 864-1440 MHzsignal. For example, referring to FIG. 7B, in a preferred embodiment,the filters 808 are not required due to the good spectralcharacteristics of the signals at that point. A similar situation mayapply to the other filters shown throughout, or the filtering may beachieved by different filters and/or filters placed in differentlocations. Similarly, amplification may be achieved by devices otherthan the various gain blocks shown. In a preferred embodiment, both RFdown-converter 624 and RF up-converter 644 do not contain variable gainelements. As one final example, in FIGS. 4-8, some functionality isimplemented in the digital domain while other functionality isimplemented in the analog domain. This apportionment between digital andanalog may be different for other implementations. Other variations willbe apparent.

The FDM aspect of preferred embodiment 400 has been described in thecontext of combining 64 low-speed channels 240 into a single opticalhigh-speed channel 120. The invention is in no way limited by thisexample. Different total numbers of channels, different data rates foreach channel, different aggregate data rate, and formats and protocolsother than the STS/OC protocol are all suitable for the currentinvention. In fact, one advantage of the FDM approach is that it iseasier to accommodate low-speed channels which use different data ratesand/or different protocols. In other words, some of the channels 240Bmay use data rate A and protocol X; while others may use data rate B andprotocol Y, while yet others may use data rate C and protocol Z. In theFDM approach, each of these may be allocated to a different carrierfrequency and they can be straightforwardly combined so long as theunderlying channels are not so wide as to cause the different carriersto overlap. In contrast, in the TDM approach, each channel is allocatedcertain time slots and, essentially, will have to be converted to a TDMsignal before being combined with the other channels.

Another advantage is lower cost. The FDM operations may be accomplishedwith low-cost components commonly found in RF communication systems.Additional cost savings are realized since the digital electronics suchas modulator 640 and demodulator 620 operate at a relatively low datarate compared to the aggregate data rate. The digital electronics needonly operate as fast as the data rate of the individual low-speedchannels 240. This is in contrast to TDM systems, which require adigital clock rate that equals the aggregate transmission rate. ForOC-192, which is the data rate equivalent to the high-speed channels 120in system 100, this usually requires the use of relatively expensivegallium arsenide integrated circuits instead of silicon.

Moving further along transmitter 210B, E/O converter 240 preferablyincludes an optical source and an external optical modulator. Examplesof optical sources include solid state lasers and semiconductor lasers.Example external optical modulators include Mach Zehnder modulators andelectro-absorptive modulators. The optical source produces an opticalcarrier, which is modulated by the electrical high-speed channel as thecarrier passes through the modulator. The electrical high-speed channelmay be predistorted in order to increase the linearity of the overallsystem. Alternatively, E/O converter 240 may be an internally modulatedlaser. In this case, the electrical high-speed channel drives the laser,the output of which will be a modulated optical beam (i.e., the opticalhigh-speed channel 120B).

The wavelength of the optical high-speed channel may be controlled usinga number of different techniques. For example, a small portion of theoptical carrier may be extracted by a fiber optic splitter, whichdiverts the signal to a wavelength locker. The wavelength lockergenerates an error signal when the wavelength of the optical carrierdeviates from the desired wavelength. The error signal is used asfeedback to adjust the optical source (e.g., adjusting the drive currentor the temperature of a laser) in order to lock the optical carrier atthe desired wavelength. Other approaches will be apparent.

The counterpart on the receiver 210A is O/E converter 220, whichtypically includes a detector such as an avalanche photo-diode orPIN-diode. In an alternate approach, O/E converter 220 includes aheterodyne detector. For example, the heterodyne detector may include alocal oscillator laser operating at or near the wavelength of theincoming optical high-speed channel 120A. The incoming opticalhigh-speed channel and the output of the local oscillator laser arecombined and the resulting signal is detected by a photodetector. Theinformation in the incoming optical high-speed channel can be recoveredfrom the output of the photodetector. One advantage of heterodynedetection is that the thermal noise of the detector can be overcome andshot noise limited performance can be obtained without the use of fiberamplifiers.

The modularity of the FDM approach also makes the overall system moreflexible and scaleable. For example, frequency bands may be allocated tocompensate for fiber characteristics. For a 70 km fiber, there istypically a null around 7 GHz. With the FDM approach, this null may beavoided simply by not allocating any of the frequency bands around thisnull to any low-speed channel 240. As a variant, each of the frequencybands may be amplified or attenuated independently of the others, forexample in order to compensate for the transmission characteristics ofthat particular frequency band.

Various design tradeoffs are inherent in the design of a specificembodiment of an FDM-based system 100 for use in a particularapplication. For example, the type of Reed Solomon encoding may bevaried or other types of forward error correction codes (or none at all)may be used, depending on the system margin requirements. As anotherexample, in one variation of QAM, the signal lattice is evenly spaced incomplex signal space but the total number of states in the QAMconstellation is a design parameter which may be varied. The optimalchoices of number of states and other design parameters formodulator/demodulator 640/620 will depend on the particular application.Furthermore, the modulation may differ on some or all of the low speedchannels. For example, some of the channels may use PSK modulation,others may use 16-QAM, others may use 4-QAM, while still others may usean arbitrary complex constellation. The choice of a specific FDMimplementation also involves a number of design tradeoffs, such as thechoices of intermediate frequencies, whether to implement components inthe digital or in the analog domain, and whether to use multiple stagesto achieve the multiplexing.

As a numerical example, in one embodiment, a (187,204) Reed-Solomonencoding may be used with a rate ¾ 16-QAM trellis code. The (187,204)Reed-Solomon encoding transforms 187 bytes of data into 204 bytes ofencoded data and the rate ¾ 16-QAM trellis code transforms 3 bits ofinformation into a single 16-QAM symbol. In this example, a singlelow-speed channel 240B, which has a base data rate of 155 Mbps wouldrequire a symbol rate of 155 Mbps×(204/187)×(⅓)=56.6 Mega symbols persecond. Including an adequate guard band, a typical frequency band wouldbe about 72 MHz to support this symbol rate. Suppose, however, that itis desired to decrease the bandwidth of each frequency band. This couldbe accomplished by changing the encoding and modulation. For example, a(188,205) Reed-Solomon code with a rate ⅚ 64-QAM trellis code wouldrequire a symbol rate of 155 Mbps×(205/188)×(⅕)=33.9 Megasymbols persecond or 43 MHz frequency bands, assuming proportional guard bands.Alternately, if 72 MHz frequency bands were retained, then the data ratecould be increased.

As another example, an optical modulator 240 with better linearity willreduce unwanted harmonics and interference, thus increasing thetransmission range of system 100. However, optical modulators withbetter linearity are also more difficult to design and to produce.Hence, the optimal linearity will depend on the particular application.An example of a system-level tradeoff is the allocation of signal powerand gain between the various components. Accordingly, many aspects ofthe invention have been described in the context of the preferredembodiment of FIGS. 3-8 but it should be understood that the inventionis not to be limited by this specific embodiment.

It should be noted that the embodiments described above are exemplaryonly and many other alternatives will be apparent. For example, in theembodiments discussed above, the low-speed channels 240 were combinedinto an electrical high-speed channel using solely frequency divisionmultiplexing. For example, each of the 64 low-speed channels 240B waseffectively placed on a carrier of a different frequency and these 64carriers were then effectively combined into a single electricalhigh-speed channel solely on the basis of different carrier frequencies.This is not meant to imply that the invention is limited solely tofrequency division multiplexing to the exclusion of all other approachesfor combining signals. In fact, in alternate embodiments, otherapproaches may be used in conjunction with frequency divisionmultiplexing. For example, in one approach, 64 low-speed channels 240Bmay be combined into a single high-speed channel 120 in two stages, onlythe second of which is based on frequency division multiplexing. Inparticular, 64 low-speed channels 240B are divided into 16 groups of 4channels each. Within each group, the 4 channels are combined into asingle signal using 16-QAM (quadrature amplitude modulation). Theresulting QAM-modulated signals are frequency-division multiplexed toform the electrical high-speed channel.

As another example, it should be clear that the tributaries 160 maythemselves be combinations of signals. For example, some or all of theOC-3/OC-12 tributaries 160 may be the result of combining several lowerdata rate signals, using either frequency division multiplexing or othertechniques. In one approach, time division multiplexing may be used tocombine several lower data rate signals into a single OC-3 signal, whichserves as a tributary 160.

As a final example, frequency division multiplexing has been used in allof the preceding examples as the method for combining the low-speedchannels 240 into a high-speed channel 120 for transmission acrossoptical fiber 104. Other approaches could also be used. For example, thelow-speed channels 240 could be combined using wavelength divisionmultiplexing, in which the combining of channels occurs in the opticaldomain rather than in the electrical domain. In this approach, thelow-speed channels are optical in form, the optical power of eachlow-speed channel is adjusted, and the power-adjusted optical low-speedchannels are combined using wavelength division multiplexing rather thanfrequency division multiplexing. Many of the principles described abovemay also be applied to the wavelength division multiplexing approach.Although the invention has been described in considerable detail withreference to certain preferred embodiments thereof, other embodimentsare possible. Therefore, the scope of the appended claims should not belimited to the description of the preferred embodiments containedherein.

1. In an optical fiber communications system including a transmitternode coupled to a receiver node by an optical fiber, a method forsynchronizing the receiver node with the transmitter node, the methodcomprising: at the transmitter node: generating a reference signal;synchronizing the transmitter node with the reference signal; whereinsaid synchronizing the transmitter node comprises adjusting the datarate of a data signal to match the rate of the reference signal;modulating the reference signal onto an optical signal in a firstfrequency band and modulating the data signal onto the optical signal ina second frequency band, the first frequency band being different fromthe second frequency band; and transmitting the optical signal acrossthe optical fiber to the receiver node; and at the receiver node:receiving the optical signal; recovering the reference signal and thedata signal from the optical signal; and synchronizing the receiver nodewith the recovered reference signal; wherein said synchronizing thereceiver node comprises adjusting the data rate of the recovered datasignal to match the rate of the recovered reference signal.
 2. Themethod of claim 1 wherein: each of the transmitter node and the receivernode includes a local oscillator; synchronizing the transmitter nodewith the reference signal comprises synchronizing a local oscillator atthe transmitter node with the reference signal; and synchronizing thereceiver node with the recovered reference signal comprisessynchronizing a local oscillator at the receiver node with the recoveredreference signal.
 3. The method of claim 1 wherein: modulating thereference signal and the data signal onto an optical signal comprises:generating a harmonic of the reference signal; and modulating theharmonic onto the optical signal; and recovering the reference signaland the data signal from the optical signal comprises: recovering theharmonic from the optical signal; and frequency dividing the harmonic torecover the reference signal.
 4. The method of claim 1 wherein:modulating the reference signal and the data signal onto an opticalsignal comprises: frequency division multiplexing the reference signalwith a plurality of electrical low-speed channels to form an electricalhigh-speed channel; and converting the electrical high-speed channelfrom electrical to optical form to form the optical signal; andrecovering the reference signal and the data signal from the opticalsignal comprises: converting the optical signal from optical toelectrical form to recover the electrical high-speed channel; andfrequency division demultiplexing the reference signal from theelectrical high-speed channel.
 5. The method of claim 4 wherein, in theelectrical high-speed channel, the reference signal is located at afrequency lower than that of the electrical low-speed channels.
 6. Themethod of claim 1 wherein: each of the transmitter node and the receivernode includes a local oscillator; synchronizing the transmitter nodewith the reference signal comprises locking a local oscillator used inthe transmitter node to the reference signal; and synchronizing thereceiver node with the recovered reference signal comprises locking alocal oscillator used in the receiver node to the recovered referencesignal.
 7. The method of claim 6 wherein: modulating the referencesignal onto an optical signal comprises: generating a harmonic of thereference signal; and modulating the harmonic onto the optical signal;and recovering the reference signal from the optical signal comprises:recovering the harmonic from the optical signal; and frequency dividingthe harmonic to recover the reference signal.
 8. The method of claim 7wherein: the substep of modulating the harmonic onto an optical signalcomprises: frequency division multiplexing the harmonic with a pluralityof electrical low-speed channels to form an electrical high-speedchannel; and converting the electrical high-speed channel fromelectrical to optical form to form the optical signal; and the substepof recovering the harmonic from the optical signal comprises: convertingthe optical signal from optical to electrical form to recover theelectrical high-speed channel; and frequency division demultiplexing theharmonic from the electrical high-speed channel.
 9. The method of claim8 wherein, in the electrical high-speed channel, the harmonic is locatedat a frequency lower than that of the electrical low-speed channels. 10.The method of claim 8 wherein each of the electrical low-speed channelsand the harmonic is allocated a different frequency band within theelectrical high-speed channel and within the optical signal.
 11. Anoptical fiber communications system for transmitting at least twolow-speed channels across the communications system, the communicationssystem comprising: a transmitter node including: a local oscillator forgenerating a reference signal; and an FDM multiplexer coupled to thelocal oscillator for combining the low-speed channels with the referencesignal into an electrical high-speed channel, wherein the referencesignal and the low speed channels are separate signals in the electricalhigh-speed channel, wherein each of the low-speed channels and thereference signal is allocated a different frequency band within theelectrical high-speed channel; and a receiver node coupled to thetransmitter node by an optical fiber, the receiver node including: anFDM demultiplexer for recovering the reference signal from theelectrical high-speed channel; a local oscillator; and electronicscoupled to the local oscillator and the FDM demultiplexer forsynchronizing the local oscillator with the recovered reference signal.12. The optical fiber communications system of claim 11 wherein: thetransmitter node further includes: electronics coupled between the localoscillator and the FDM multiplexer for generating a pilot tone from aharmonic of the reference signal, wherein the FDM multiplexer combinesthe low-speed channels with the pilot tone into an electrical high-speedchannel; and an E/O converter coupled to the FDM multiplexer forconverting the electrical high-speed channel into an optical high-speedchannel; and the receiver node further includes: an O/E convertercoupled to the FDM demultiplexer for receiving the optical high-speedchannel and converting it to the electrical high-speed channel; whereinthe FDM demultiplexer recovers the pilot tone from the electricalhigh-speed channel and the electronics in the receiver node recover thereference signal from the pilot tone.
 13. The optical fibercommunications system of claim 12 wherein, in the electrical high-speedchannel, the pilot tone is located at a frequency lower than that of theelectrical low-speed channels.
 14. The optical fiber communicationssystem of claim 12 wherein each of the electrical low-speed channels andthe pilot tone is allocated a different frequency band within theelectrical high-speed channel and within the optical high-speed channel.15. A transmitter node for transmitting at least two low-speed channelsacross an optical fiber communications system, wherein the transmitternode comprises: a generator configured to generate a reference signal; asynchronizer configured to adjust a data rate of a data signal to matcha rate of the reference signal; a modulator configured to modulate thereference signal onto an optical signal in a first frequency band andmodulate the data signal onto the optical signal in a second frequencyband, the first frequency band being different from the second frequencyband; and circuitry configured to transmit the optical signal across theoptical fiber.
 16. The transmitter node of claim 15, wherein thegenerator is further configured to generate a harmonic of the referencesignal; and wherein to modulate the reference signal onto the opticalsignal, the modulator is further configured to modulate the harmoniconto the optical signal.
 17. The transmitter node of claim 15, whereinthe modulator includes an FDM multiplexer; wherein to modulate thereference signal and the data signal onto an optical signal, the FDMmultiplexer is configured to frequency division multiplexing thereference signal with a plurality of electrical low-speed channels toform an electrical high-speed channel; and wherein an electrical tooptical (L/O) converter is configured to convert the electricalhigh-speed channel from electrical to optical form to form the opticalsignal.
 18. The transmitter node of claim 17 wherein, within theelectrical high-speed channel, the reference signal is located at afrequency lower than that of the electrical low-speed channels.
 19. Thetransmitter node of claim 15, wherein the transmitter node includes alocal oscillator; and wherein the transmitter node is further configuredto lock the local oscillator to the reference signal.
 20. Thetransmitter node of claim 19, wherein the transmitter node is furtherconfigured to generate a pilot tone from a harmonic of the referencesignal; and wherein to modulate the reference signal onto the opticalsignal, the modulator is further configured to modulate the pilot toneonto the optical signal.
 21. The transmitter node of claim 20, whereinto modulate the pilot tone onto an optical signal, the FDM multiplexeris configured to frequency division multiplex the pilot tone with aplurality of electrical low-speed channels to form an electricalhigh-speed channel.
 22. The transmitter node of claim 21 wherein, withinthe electrical high-speed channel, the pilot tone is located at afrequency lower than that of the electrical low-speed channels.
 23. Thetransmitter node of claim 21 wherein each of the electrical low-speedchannels and the pilot tone is allocated a different frequency bandwithin the electrical high-speed channel and within the optical signal.24. A receiver node for receiving at least two low-speed channelstransmitted across an optical fiber communications system, wherein thereceiver node comprises: circuitry configured to receive an opticalsignal; a demodulator configured to recover a reference signal and adata signal from the optical signal, wherein the reference signal ismodulated onto the optical signal in a first frequency band and the datasignal is modulated onto the optical signal in a second frequency band,the first frequency band being different from the second frequency band;and a synchronizer configured to adjust a data rate of the recovereddata signal to match a rate of the recovered reference signal.
 25. Thereceiver node of claim 24, wherein to recover the reference signal fromthe optical signal, the demodulator is further configured to: recover aharmonic of the reference signal from the optical signal; and frequencydivide the harmonic to recover the reference signal.
 26. The receivernode of claim 24, wherein the receiver node includes an electrical tooptical (O/L) converter coupled to an FDM demultiplexer; wherein the O/Econverter is configured to convert the optical signal from optical toelectrical form to recover the electrical high-speed channel; andwherein the FDM demultiplexer is configured to frequency divisiondemultiplexing the reference signal from the electrical high-speedchannel.
 27. The receiver node of claim 26 wherein, in the electricalhigh-speed channel, the reference signal is located at a frequency lowerthan that of the electrical low-speed channels.
 28. The receiver node ofclaim 24, wherein the receiver node is further configured to lock thesecond local oscillator to the recovered reference signal.
 29. Thereceiver node of claim 28, wherein to recover the reference signal fromthe optical signal, the demodulator is further configured to: recover apilot tone from the optical signal; and frequency divide the pilot toneto recover the reference signal.
 30. The receiver node of claim 29,wherein to recover the pilot tone from the optical signal, the FDMdemultiplexer is configured to frequency division demultiplex the pilottone from the electrical high-speed channel.
 31. The receiver node ofclaim 30 wherein, in the electrical high-speed channel, the pilot toneis located at a frequency lower than that of the electrical low-speedchannels.
 32. The receiver node of claim 30 wherein each of theelectrical low-speed channels and the pilot tone is allocated adifferent frequency band within the electrical high-speed channel andwithin the optical high-speed channel.
 33. An optical fibercommunications system for transmitting at least two low-speed channelsacross the communications system, the communications system comprising:a transmitter node configured to: generate a reference signal; modulatethe reference signal onto an optical signal in a first frequency bandand modulate the data signal onto the optical signal in a secondfrequency band, the first frequency band being different from the secondfrequency band; and transmit the optical signal across the opticalfiber; and a receiver node configured to: receive the optical signal;recover the reference signal and the data signal from the opticalsignal; and adjust a data rate of the recovered data signal to match arate of the recovered reference signal.
 34. The optical fibercommunications system of claim 33, wherein the transmitter node isfurther configured to adjust the data rate of a data signal to match therate of the reference signal.
 35. The optical fiber communicationssystem of claim 33, wherein to modulate the reference signal onto anoptical signal, the transmitter node is further configured to: generatea harmonic of the reference signal; and modulate the harmonic onto theoptical signal.
 36. The optical fiber communications system of claim 35,wherein the receiver node is further configured to: recover the harmonicfrom the optical signal; and frequency divide the harmonic to recoverthe reference signal.
 37. The optical fiber communications system ofclaim 33, wherein the transmitter node includes an FDM multiplexercoupled to an electrical to optical (L/O) converter; wherein to modulatethe reference signal and the data signal onto an optical signal, the FDMmultiplexer is configured to frequency division multiplexing thereference signal with a plurality of electrical low-speed channels toform an electrical high-speed channel; and wherein the E/O converter isconfigured to convert the electrical high-speed channel from electricalto optical form to form the optical signal.
 38. The optical fibercommunications system of claim 37, wherein the receiver node includes anO/E converter coupled to an FDM demultiplexer; wherein the O/E converteris configured to convert the optical signal from optical to electricalform to recover the electrical high-speed channel; and wherein the FDMdemultiplexer is configured to frequency division demultiplexing thereference signal from the electrical high-speed channel.
 39. The opticalfiber communications system of claim 38 wherein, in the electricalhigh-speed channel, the reference signal is located at a frequency lowerthan that of the electrical low-speed channels.
 40. The optical fibercommunications system of claim 33, wherein the transmitter node includesa first local oscillator; and wherein the transmitter node is furtherconfigured to lock the first local oscillator to the reference signal.41. The optical fiber communications system of claim 40, wherein tomodulate the reference signal onto an optical signal, the transmitternode is further configured to: generate a pilot tone from a harmonic ofthe reference signal; and modulate the pilot tone onto the opticalsignal.
 42. The optical fiber communications system of claim 41 whereinto modulate the pilot tone onto an optical signal, the FDM multiplexeris configured to frequency division multiplex the pilot tone with aplurality of electrical low-speed channels to form an electricalhigh-speed channel.
 43. The optical fiber communications system of claim41, wherein the receiver node is further configured to: recover thepilot tone from the optical signal; and frequency divide the pilot toneto recover the reference signal.
 44. The optical fiber communicationssystem of claim 33, wherein the receiver node includes a second localoscillator; and wherein the receiver node is further configured to lockthe second local oscillator to the recovered reference signal.
 45. Theoptical fiber communications system of claim 44, wherein to recover thepilot tone from the optical signal, the FDM demultiplexer is configuredto frequency division demultiplex the pilot tone from the electricalhigh-speed channel.
 46. In an optical fiber communications systemincluding a transmitter node coupled to a receiver node by an opticalfiber, a method for synchronizing the receiver node with the transmitternode, the method comprising: at the transmitter node: generating areference signal; modulating the reference signal onto an optical signalin a first frequency band and modulating the data signal onto theoptical signal in a second frequency band, the first frequency bandbeing different from the second frequency band; and transmitting theoptical signal across the optical fiber; and at the receiver node:receiving the optical signal; recovering the reference signal and thedata signal from the optical signal; and adjusting a data rate of therecovered data signal to match a rate of the recovered reference signal.47. The method of claim 46, further comprising adjusting a data rate ofa data signal at the transmitter node to match a rate of the referencesignal.
 48. The method of claim 46, wherein modulating the referencesignal and the data signal onto an optical signal comprises: generatinga harmonic of the reference signal; and modulating the harmonic onto theoptical signal.
 49. The method of claim 48, further comprising:recovering the harmonic from the optical signal at the receiver node;and frequency dividing the harmonic to recover the reference signal. 50.The method of claim 46, further comprising: frequency divisionmultiplexing the reference signal with a plurality of electricallow-speed channels to form an electrical high-speed channel at thetransmitter node; and converting the electrical high-speed channel fromelectrical to optical form to form the optical signal.
 51. The method ofclaim 50, further comprising: converting the optical signal from opticalto electrical form at the receiver node to recover the electricalhigh-speed channel; and frequency division demultiplexing the referencesignal from the electrical high-speed channel.
 52. The method of claim51 wherein in the electrical high-speed channel, the reference signal islocated at a frequency lower than that of the electrical low-speedchannels.
 53. The method of claim 46, wherein the transmitter nodeincludes a first local oscillator, the method further comprising lockingthe first local oscillator to the reference signal.
 54. The method ofclaim 53, wherein modulating the reference signal and the data signalonto an optical signal comprises: generating a pilot tone from aharmonic of the reference signal at the transmitter node; and modulatingthe pilot tone onto the optical signal.
 55. The method of claim 54wherein modulating the pilot tone onto an optical signal furthercomprises frequency division multiplexing the pilot tone with aplurality of electrical low-speed channels to form an electricalhigh-speed channel.
 56. The method of claim 54, further comprising:recovering the pilot tone from the optical signal; and frequencydividing the pilot tone to recover the reference signal.
 57. The methodof claim 56, wherein the receiver node includes a second localoscillator, the method further comprising locking the second localoscillator to the recovered reference signal.
 58. The method of claim56, wherein recovering the pilot tone from the optical signal furthercomprises frequency division demultiplexing the pilot tone from theelectrical high-speed channel.
 59. A method for transmitting at leasttwo low-speed channels across an optical fiber communications system,the method comprising: generating a reference signal; adjusting a datarate of a data signal to match a rate of the reference signal;modulating the reference signal onto an optical signal in a firstfrequency band and modulating the data signal onto the optical signal ina second frequency band, the first frequency band being different fromthe second frequency band; and transmitting the optical signal acrossthe optical fiber.
 60. The method of claim 59, further comprising:generating a harmonic of the reference signal; and modulating theharmonic onto the optical signal.
 61. The method of claim 59, furthercomprising: frequency division multiplexing the reference signal with aplurality of electrical low-speed channels to form an electricalhigh-speed channel; and converting the electrical high-speed channelfrom electrical to optical form to form the optical signal.
 62. Themethod of claim 61 wherein, within the electrical high-speed channel,the reference signal is located at a frequency lower than that of theelectrical low-speed channels.
 63. The method of claim 59, furthercomprising locking a local oscillator to the reference signal.
 64. Themethod of claim 63, further comprising: generating a pilot tone from aharmonic of the reference signal; and modulating the pilot tone onto theoptical signal.
 65. The method of claim 64, wherein to modulate thepilot tone onto an optical signal, the method comprises frequencydivision multiplexing the pilot tone with a plurality of electricallow-speed channels to form an electrical high-speed channel.
 66. Themethod of claim 65 wherein, within the electrical high-speed channel,the pilot tone is located at a frequency lower than that of theelectrical low-speed channels.
 67. A method for receiving at least twolow-speed channels transmitted across an optical fiber communicationssystem, the method comprising: receiving the optical signal; recoveringa reference signal and a data signal from the optical signal, whereinthe reference signal is modulated onto the optical signal in a firstfrequency band and the data signal is modulated onto the optical signalin a second frequency band, the first frequency band being differentfrom the second frequency band; and adjusting a data rate of therecovered data signal to match a rate of the recovered reference signal.68. The method of claim 67, wherein recovering the reference signal fromthe optical signal further comprises: recovering a harmonic of thereference signal from the optical signal; and frequency dividing theharmonic.
 69. The method of claim 67, further comprising: converting theoptical signal from optical to electrical form to recover the electricalhigh-speed channel; and frequency division demultiplexing the referencesignal from the electrical high-speed channel.
 70. The method of claim69 wherein, in the electrical high-speed channel, the reference signalis located at a frequency lower than that of the electrical low-speedchannels.
 71. The method of claim 67, further comprising locking a localoscillator to the recovered reference signal.
 72. The method of claim71, wherein to recover the reference signal, the method furthercomprises: recovering a pilot tone from the optical signal; andfrequency dividing the pilot tone.
 73. The method of claim 72, whereinto recover the pilot tone from the optical signal, the method comprisesfrequency division demultiplexing the pilot tone from the electricalhigh-speed channel.
 74. The method of claim 73 wherein, in theelectrical high-speed channel, the pilot tone is located at a frequencylower than that of the electrical low-speed channels.
 75. The method ofclaim 73 wherein each of the electrical low-speed channels and the pilottone is allocated a different frequency band within the electricalhigh-speed channel and within the optical high-speed channel.